phần 4

Silicon micromachined infrared sensor with tunable wavelength
selectivity for application in infrared spectroscopy

Lehrstuhl für Technische Elektronik, Technische Universität München, Arcisstraße 21, D-80333 Munich, Germany

Abstract

The techniques of silicon micromaching have been used to develop a miniature infrared sensor with tunable wavelength selectivity for application in infrared spectroscopy. The infrared sensor consists of a tunable interference filter in front of a wide-band detector. The applicable spectral bandwidth ranges from 1.5 to 7.5 qjn. The resolution is better than 25 nm over the whole range. The wavelength tuning and parallelism control of the mirrors is carried out by electrostatic forces, varying the voltage at the integrated disk capacitors. The transmitted infrared radiation is absorbed in a black gold layer, the rising temperature being measured by a thermopile consisting of 80 Si-Ni thermocouples. This device is expected to find application as an emission monitor for liquids and gases.

Keywords: Infrared sensors; Silicon micromachining

$Text Box: , , 4R . J 2'trndm cos ß 1 + r- sin (1-P2) \ A$

1. Introduction

Spectral analysis in the infrared region is a well- known, common and powerful method in physics and chemistry for analysing the composition of many substances [1]. For this application a silicon-based infrared sensor would be a reasonable and handy alternative to conventional infrared spectrometers, which normally tend to be large expensive devices that are difficult to tune electronically. It could be used in portable equipment for outdoor chemical analyses.

In the last few years, many powerful miniaturized spectrometers have been developed for the spectral range 300-1100 nm where silicon CCD lines can be used as detectors. The infrared sensor presented in this paper works in the spectral range where silicon is transparent. Therefore this device extends the range of handy and reasonable miniaturized spectrometers into the infrared region, which is a very interesting one for chemical analyses [1].

2. Structure and device fabrication

2,1. Tunable interference filter

A Fabry-Perot interferometer (FPI) is an optical element made up of two partially reflective mirrors

0924-4247/95/S09.50 © 1995 Elsevier Science S.A. All rights reserved SSDI 0924-4247194100932-8
separated by a distance d_m. They have to be adjusted in such a way that they are parallel to a very high degree of accuracy. The parallelism is normally achieved by a spacer ring between the mirrors separating them at the desired distance. Such an arrangement of two plane mirrors and a ring as spacer is named a Fabry- Perot etalon. The outer surfaces of the mirrors are usually coated with antireflection layers. The mirror itself is often built up by dielectric layers to obtain high reflectance and to minimize absorbance.

Such a set-up transmits radiation according to the following equation:

T^z

(i-R²) where I = transmitted intensity, R = reflection coefficient of the reflecting layer, T=transmission coefficient of the reflecting layer, ifi=change of phase on reflection at the reflecting layer, /3 = angle of incidence of the beam, n = refractive index of the spacer, d_m = thickness of the space and A=wavelength of radiation.

The reflecting layers are assumed to be identical on both mirrors. This transmission characteristic consists of a number of very sharp transmission peaks, which are caused by multiple reflections of the radiation in the Fabry-Perot etalon [3].

Whenever a multiple number of half wavelengths matches between the two mirrors, radiation will pass through:

d_m = iy (/ = 1,2,3...) (2)

=>A₀ = 2^ (/ = 1,2,3...) (3)

The distance between two transmission peaks, AA₀, is therefore

-1+?)

+!>■) ⁼ W

The half-width HW of a transmission peak is given by (¿ = 1,2,3...) (5)

The finesse Q can be calculated by

A₀ _ iirR^mHW~,₀ ⁼ I

(see Fig. 1).

Using highly reflecting mirrors, a small variation of the mirror separation d_m causes a significant change of the transmission characteristic. Therefore the realization of a miniaturized Fabry-Perot interferometer is possible [4].

The integrated Fabry-Perot interferometer consists of two silicon parts placed on each other. A cross

^-0

Fig. 1. Optical characteristics at the transmission curve of a Fabry- Perot interferometer.

section of the whole infrared sensor is shown in Fig. 2. The overall dimensions are 20 mmX20 mm X 0.8 mm. On both parts high reflective mirrors are applied by sputtering processes. The reflective index of the mirrors is about 0.92. To reduce substantial Fresnel reflection losses at the outer silicon surface (rz_Si = 3.4) antireflection coatings have been applied.

The first part has a thin membrane with the dimensions 10 mmXlO mmXl3 ¡im at its centre, which is fabricated by anisotropic wet etching. In the second part a hollow is structured through dry etching methods. The depth of the hollow determines the distance between the two Fabry-Perot mirrors. The roughness of the hollow surface is less than 5 nm. Around the mirror on the membrane four capacitor plates are placed electrically isolated from each other. On the other part a metallic layer of the same size is applied. The two wafers are bonded together to form the optical etalon. So there are four disk capacitors to control the wavelength selectivity by applying a voltage to them. The electrostatic forces thus produced pull the membrane forward to the opposite surface, reducing the distance between the two mirrors. The reaction force is given by the elastic restoring force of the silicon membrane. With four disk capacitors it is not only possible to control the mirror spacing but also the mirror parallelism to achieve maximum finesse of the system. The capacity is given by

C=^-<f₀e_r (7)

Therefore the value of the capacity changes by variation of the mirror distance. This effect is used to stabilize the mirror distance in an active feedback loop, to

Fig. 2. Cross section of the tunable infrared sensor.

$Text Box: 2TT\ L$ Text Box:
Fig. 3. Schematic layout of the mirror-distance controller.

minimize environmental stimulation (microphonia) [2] and to balance manufacturing irregularities. The schematic layout is shown in Fig. 3. It consists of an oscillator generating frequency steps with fixed (but adjustable) distances. The distance will determine the resolution and the absolute frequency will determine the wavelength. This frequency is the reference for the automatic control system, consisting of four phase-lock loops (PLLs).

The capacities C_1A3A sensor ^are the frequency-determining elements of four LC oscillators. On changing the value of the capacitors, the related frequency changes according to the equation

^/à-⁴ 2tt(LC₁_₄)^1/2 (see Fig. 4). Using the PLL principle for each capacitor, it is possible to regulate the mirror distance d_m( = plate distance of the capacitors) in such a way that no influence of microphonia can be determined. fjd_m ranges from about 4 to about 15 kHz nm'¹.

2.2. Wide-band infrared detector

This part of the infrared sensor consists of two functional groups: an infrared absorber and a thermal detector.

The infrared absorber is placed at the backside of the silicon membrane upon the wet etched surface. It consists of black gold deposited in a rather poor vacuum with high evaporation power. It converts the transmitted radiation wavelength unselectively into heat, which can be measured with a thermal detector [5]. The thermal detector is a thermopile composed of 80 Si—Ni thermocouples in series. The hot ends of the thermocouples are arranged under the capacitor plates and around the mirror, whereas the cold junctions are placed at the bulk material. The surfaces of the bond pads of the thermopile are of the same material and set so closely together that no further thermovoltage occurs. The resulting thermovoltage is about 37 mV K^_1 at room temperature and the time constant is about 30 ms. The estimated noise equivalent power is about

^FThe^cpne^l^XlO^WHz-^²

and the responsivity is about

^Thermopi.e=H0 VW^_1

3. System set-up for infrared spectroscopy

A conventional set-up for infrared spectroscopy is shown in Fig. 5. It consists of an infrared source (e.g., a silicon planar pellistor), the medium of the investigated and the infrared sensor. The significant advantage of this infrared sensor is the outer dimensions, which are very small compared to those of conventional infrared spectrometers. To achieve a small and compact set-up it is useful to have short absorption lengths. Because of their high absorption coefficients, fluid and more concentrated gases need only a short length for high

IR-source medium IR-senso^p

absorption. Therefore the main application area will be the control of fluids and the emission control of gas outlets. In such a set-up it is possible to have no lenses or mirrors. The cuvette for the medium to be investigated is a simple infrared transmitting tube. But it is also possible to use multipass infrared cuvettes to detect low-concentration gases or solvents by giving up the advantage of having a small measurement set-up.

4. Conclusions

A new silicon micromachined infrared sensor has been presented for use in infrared spectroscopy with tunable wavelength selectivity. Using a Fabry-Perot interformeter offers the unique advantage of having a linewidth which can be applied to the measurement requirements by using a suitable mirror spacing and finesse. The big advantages of this new device are its small dimensions and low cost compared with standard infrared spectrometers. This allows reasonable and handy infrared spectrometers to be built for outdoor use. The limit in wavelength of 1100 nm for miniaturized spectrometers could be extended up to about 7.5 pm. It might be of interest that the structure of the miniaturized Fabry-Pdrot interferometer can also be used as a voltage-controlled oscillator with a high tunable range.

Acknowledgement

The author wishes to thank Deutsche Forschungsgemeinschaft (DFG) for financial support.

References

[1] B.P.Straughan,Specfroicc/jy, Vol. 2, Chapman and Hall, London, 1976, pp. 138-265.

[2] W. Albertshofcr, A tunable ‘spcctrometerdiode’ with a spectral resolution of 3 nm in the 660-900 nm range, Sensors and Actuators A, 25-27 (1991) 443-447.

[3] J.M. Vaughan, The Fabry-Tirol Interferometer, Adam Hilger, Bristol, 1989, pp. 89-177.

[4] P. Hariharan, Optical Interferometry, Academic Press, Sydney, 1985, pp. 79-93.

[5] R.H. Kingston, Detection of Optical and Infrared Radiation, Springer, Berlin, 1978, pp. 83-100.

A Low Cost Monolithic Accelerometer

S. J. Sherman, W. K. Tsang, T. A. Core, D. E. Quinn
Analog Devices Semiconductor
Wilmington, MA 01887

1, INTRODUCTION

The ADXL50 is a complete scaled and temperature compensated surface micro-machined accelerometer with an output voltage proportional to acceleration. Full scale measurement range is ±50g, with unpowered shock survival at 2000g. Ultimately, signal span accuracy of 5% should be possible for a temperature range of -55°C to +125°C and a supply range of 5V ±0.25 V. Bandwidth up to 1.5KHz is programmable with a single external capacitor.

A digitally activated self-test will electrostatically deflect a functional beam so that a -50g acceleration is indicated.

An uncommitted amplifier, with rail-to-rail output range, and a reference allow re-scaling and offsetting of the raw output signal (1.8V ±1.0V at ±50g). Capacitors can be introduced in the gain network surrounding the uncommitted amp so that 2 poles of low pass filtering are possible without the addition of off-chip active circuitry.

The ADXL50's objective specifications were crafted for crash detection in second generation automotive air bag systems which rely on single point sensing and per model programmable crash signature analysis for dramatic system cost reduction.

2. TECHNOLOGY BASE

The sensor's low cost objective, ultimately S5 in automotive volumes, dictates a technology base that includes;

1. a monolithic approach, with integrated sensor and BiMOS interface circuitry

2. small chip size, 120x120 mil²

3. utilization of familiar materials and production processes

4. the simplest possible mechanical structure, a single layer of self-supporting patterned polysilicon above the substrate surface

5. standard packaging

6. exploitation of established technique, laser wafer trimmed (LWT) thin film resistors, for achieving performance objectives

3. SENSOR GEOMETRY

Figure 1. is a depiction of the sensor's essential functional elements, which are formed from a single layer of patterned polysilicon (processed on a layer of sacrificial oxide 1.6 um thick). The elements stand on the substrate at "anchor" points, a result of pre-pattemed holes in the sacrificial oxide. The sensor, a differential capacitor, exists in a "moat” area, roughly 600um x 400um, with interconnections from the beam elements to points external to the moat accomplished by N-i- emitter diffusions.

The large (by IC standards) nominal lateral capacitor

34 #1992 Symposium on VLSI Circuits Digest of Technical Papers
gaps, 1.3um, between the outer plates and the common center plate, and the low permittivity of dry nitrogen, necessitate the paralleling of 42 unit cells to achieve 0.1 pf for each side of the differential capacitor. At that sensor source impedance level adequate signal-to-noise performance is possible.

4. SYSTEM BLOCK DIAGRAM AND SENSITIVITY EQUATION

The sensor beam is electrostatically force-balanced so that the inertial force, Fi =ma, is primarily balanced by a net electrostatic force, F_E, created by a change in the beam voltage. As will be explained, this beam voltage change, AV₀, is linearly related to acceleration, a, with the sensitivity being

A V₀ _ md*

a 2Ap6oVr(1 + 1/T) (i)

where do = capacitor gap m = beam mass A_P = plate area T = loop gain Eq = permittivty of nitrogen V_R = 1/2 DC voltage difference between the outer plates

Figure 2. is a simplified system diagram representing the essential elements in a forced-balanced scheme. Complementary 1MHz square waves, centered around V_Rand -V_R are applied to the outer plates of the sensor. The low input capacitance buffer is to prevent loading of the sensor. The synchronous demodulator detects and amplifies the 1MHz beam node signal proportional to beam deflection. The low pass filter removes 2MHz spiking, a result of the demodulation process, and sets a dominant loop pole for overall frequency compensation.

Two concurrent processes exist at the beam node;

1. position sensing, at 1MHz. For a translating center plate and fixed outer plates, an ideal parallel plate treatment reveals that output per unit deflection is first order linear, i.e.

V^ = V_P x/do (2)

where V_P is peak carrier amplitude and x is deflection from center,

2. force projection on the beam, accomplished by a nonzero value of V₀ applied to the beam through the 3 megohm resistor (R). The large value of resistance prevents the 1MHz signal, sourced by only the 0.2pf, from being reduced through loading.

The 1MHz beam node signal is a classical error signal which is driven to zero by the global negative feedback

Text Box: -55>T>125°C 4.75>VS>5.25V -55>T>125 c 4.75>VS>5.25V BW = lKHz

loop, which adjusts V₀ to create the net electrostatic force balancing the inertial force, with equilibrium at x = 0, For a two parallel plates, the attractive electrostatic force is

F = e₀ A_PV²/2d² (3)

For the beam, the net force is the sum of attractive forces to each of the outer plates,

F_e = 2A_P£₀V_RV₀/d₀² (4)

If the outer plates are biased at V_R and - V_R, the center plate (beam)is biased at V₀, and the beam remains centered, X = 0. Then

F, = F_e

ma = 2Ap£_oV_RV₀/d_o²

Vo/a = mdo^ApCoVR (5)

Variables appearing in equation (5) are temperature stable in a 5% accuracy context. (V_R is slaved from a 10ppm/°C reference.)

For finite loop gain, T, the sensitivity takes the form of equation (1), with a 1 + 1/T term in the denominator. The DC loop gain, T₀, is, in fact, trimmed to a value of 10, yielding a predictable bandwidth and adequate temperature desensitization of factors in the expression for T₀, such as carrier amplitude.

Figure 3, is a more detailed block diagram representative of the chip organization.

The carrier generator, a resistively loaded differential pair of bipolars, provides complementary 1MHz square waves which are AC coupled through 50pf capacitors to the inputs of the sensor. DC plate voltages (3.4V and 0.2V)

are set with 200K resistors. The pre-amp is a low accuracy space efficient instrumentation amplifier. The self-test current, I_sx, is routed into R_ST. In the absence of acceleration the loop output V₀ will adjust so the beam node is at 1.8V, F_E = 0, and x = 0. At that condition

V₀ a 1st R-st/O + 1/T) (6)

Loop gain is trimmed at R_P1. A wafer level full scale acceleration trim technique under development leads to a calculated change in beam voltage required to force balance 50g full scale acceleration. With this calculated value, Rp2 can be trimmed so that a IV change is observed at V₀ for a 50g input.

5. EXPERIMENTAL RESULTS

Typical measured performance for the ADXL50, observed at the pre-amp output,follows. (Full scale output, F.S.O., is defined as lOOg, or ±50g, with a corresponding 2V change.)

sensitivity drift, 3.0% sensitivity PSRR, 60dB zero - g drift, lOOmV zero - g PSRR, 48dB noise, p-p, 1% F.S.O. transverse sensitivity, 2% shock survival

2000g, lOOpsec >1600g, 500psec

Photo 1 is a comparison of outputs from a shaker reference accelerometer (top) and the ADXL50 (bottom) for 20g, 100Hz excitation.